Incompletely compensated wireless power transfer system

ABSTRACT

Disclosed in the present application is an incompletely compensated wireless power transfer system. On the basis of an SS compensated wireless power transfer system topology, a primary side capacitor C 1  and a secondary side capacitor C 2  take specific values, and in combination with a phase shift frequency modulation control method, a coupling range of a system output rated power is improved. According to the present application, by finding an appropriate combination of the primary side capacitor C 1  and the secondary side capacitor C 2 , the system can output a required rated power in a larger coupling range under the condition of incomplete compensation.

TECHNICAL FIELD

The application belongs to the technical field of wireless power transmission, and relates to a wireless power transfer system, in particular to an incompletely compensated wireless power transfer system.

BACKGROUND

FIG. 1 is a simplified circuit model of a SS compensation (series-series, that is, the primary and secondary sides are series-compensated) wireless power transfer system. On the primary and secondary sides, the coil forms LC resonance with a capacitor in series connection, which is called a harmonic oscillator. The steady-state operation of this system can be described by the equations (1) (2). (R ₁ +jX ₁)I ₁ +jωM ₁₂ I ₂ =V ₁  (1) jωM ₁₂ I ₁+(R ₂ +R _(L) +jX ₂)I ₂=0  (2) where R_(i), i=1 or 2, is the parasitic resistance in the circuit, generally including the resistance of a coil and the equivalent series resistance of the capacitor; X_(i)=ωL_(i)−1/(ωC_(i)), i=1 or 2, is the reactance of the harmonic oscillator i; I₁ and I₂ are the current phasors of the primary and secondary side loops respectively; V₁ is the phasor of an input voltage; M₁₂ is the mutual inductance of the two coils; R_(L) is equivalent load resistance; ω=2πf is angular frequency (f is the frequency of the power supply).

According to the theory of wireless power transmission, X_(i)=0 (that is, full compensation) can obtain the highest power transmission efficiency and the smallest input apparent power (that is, the input voltage is in phase with the input current).

Under the premise of complete compensation, the reflected impedance produced by the secondary side in the primary side is:

$\begin{matrix} {Z_{r} = {R_{r} = \frac{\omega^{2}M_{12}^{2}}{R_{2} + R_{L}}}} & (3) \end{matrix}$

In the case of a given input voltage and equivalent load resistance, ignoring the effect of parasitic resistance, the output power of the system is:

$\begin{matrix} {P_{out} = {\frac{V_{1}^{2}}{Z_{r}} = \frac{V_{1}^{2}R_{L}}{\omega^{2}M_{12}^{2}}}} & (4) \end{matrix}$

It can be seen that the output power of the system is inversely proportional to the square of the mutual inductance.

Therefore, under the condition of complete compensation and constant operating frequency, the system can only achieve the required rated power at a certain mutual inductance value.

To illustrate with specific examples, the used coil parameters are listed in Table 1. The direct current input voltage of the system is 400V, the rated output power is 3.3 kW, and the direct current output voltage is 400V. FIG. 2 is a finite element simulation model. FIG. 3 is the curve of the output power of the system changing with the air gap distance. It can be seen that when the air gap distance is 18 cm, the system can output 3.3 kW. As the distance decreases, the output power will not reach the required power rating; when the distance increases, the output power will exceed the rated value. When the distance is greater than 18 cm, through the phase shift control of the inverter, the input voltage acting on the primary side harmonic oscillator can be reduced, but the input current at the rated output power will be greater than the value at 18 cm. For example, at 18 cm, if the input direct current voltage is 400V, after full-bridge rectification, a high-frequency square wave is obtained, and the effective value of the fundamental wave is about 360V. At this time, the input current is about 9.2 A and the output power is 3.3 kW; while at 20 cm, through the phase shift control, the effective value of the fundamental wave is reduced to 267 V, the input current is 12.4 A, and the output power is 3.3 kW, but the input current is increased by 35% compared with 18 cm.

TABLE 1 Coil parameters A plurality Diameter of each wire Number of 32 of wires of 0.1 mm; 500 wires winding winding turns Winding size 350 mm × 350 mm × Turn pitch  0 of the coil 3.2 mm Distance 1.5 mm Distance 15 mm between the between winding and magnetic the magnetic core and core aluminum plate Aluminum 555 mm × 555 mm × Core 370 mm × 370 mm × plate size 4 mm thickness 3.5 mm

It can be seen that the coupling range of the output rated power of the traditional fully compensated SS wireless power transfer system is very limited. And although the available coupling range can be increased through phase shift control, the current caps of the coils, compensation capacitors, and inverters must also be increased. Obviously, the fully compensated SS wireless power transfer system has great limitations in practical applications.

SUMMARY OF THE APPLICATION

The purpose of the present application is to provide an incompletely compensated wireless power transfer system that can substantially improve the output rated power coupling range of the system in response to the shortcomings of the prior art. The system uses an incompletely compensated solution, and the core lies in how to obtain an appropriate capacitance combination.

The technical solution adopted by the present application is as follows:

An incompletely compensated wireless power transfer system is based on the SS-compensated wireless power transfer system topology, a primary side capacitance C₁ and a secondary side capacitance C₂ take specific values, and a phase shift frequency modulation control method is combined, to increase the coupling range of the rated output power of the system;

with incomplete compensation, that is, Xi≠0, equation (3) becomes

$\begin{matrix} {Z_{r} = {R_{r} = \frac{\omega^{2}M_{12}^{2}}{R_{2} + R_{L} + {jX}_{2}}}} & (5) \end{matrix}$ ignoring the parasitic resistance, the input impedance of the system is:

$\begin{matrix} {Z_{in} = {{{jX}_{1} + Z_{r}} = {{{jX}_{1} + \frac{\omega^{2}M_{12}^{2}}{R_{L} + {jX}_{2}}} = {{\frac{\omega^{2}M_{12}^{2}}{R_{L}^{2} + X_{2}^{2}}R_{L}} + {j\left( {X_{1} - {\frac{\omega^{2}M_{12}^{2}}{R_{L}^{2} + X_{2}^{2}}X_{2}}} \right)}}}}} & (6) \end{matrix}$ assuming that the imaginary part of the input impedance can be equal to zero, the output power is:

$\begin{matrix} {P_{out} = {\frac{V_{1}^{2}}{\frac{\omega^{2}M_{12}^{2}}{R_{L}^{2} + X_{2}^{2}}R_{L}} = {\frac{V_{1}^{2}}{\omega^{2}M_{12}^{2}}\left( {R_{L} + \frac{X_{2}^{2}}{R_{L}}} \right)}}} & (7) \end{matrix}$

By comparing equations (7) and (4), it can be found that by introducing X₂, the output power of the system can be increased. For the rated output power required by the system, from equation (7), the required X₂, under any mutual inductance can be solved, and the required C₂ can be solved correspondingly. After obtaining X₂, the imaginary part of the equation (6) is set to zero, and the required X₁ can be solved, and the required C₁ can be solved accordingly. This is a theoretical calculation of a certain mutual inductance value, which can ensure the required power output and make the input voltage and input current in phase, so as to minimize the input apparent power.

In practical applications, on the one hand, V₁ can be turned down by phase shift control, so X₂ in equation (7) can have different solutions at different V₁; on the other hand, the input voltage and input current are not necessarily in phase. In a full-bridge inverter, the input current is generally required to lag behind the voltage to achieve zero-voltage turn-on.

Therefore, for an application scenario with a given coupling range and a given restriction, it is necessary to find suitable primary side capacitance C₁ and secondary side capacitance C₂, so that the system outputs the required power within this coupling range and restriction.

The primary side capacitance C₁ and the secondary side capacitance C₂ are selected as follows:

Setting restrictions:

-   1) Rated output voltage V_(out) and power P_(out); -   2) Range of operating frequency f_(min)≤f≤f_(max); -   3) Coupling range of the coil M_(12min)≤M₁₂≤M_(12max); -   4) direct current input voltage V_(DC); for example, 400V output     from a previous stage PFC. Alternating current voltage that actually     acts on the primary harmonic oscillator, which can be adjusted by     phase shift control. -   5) Maximum primary current I_(1max); affects the selection of     inverter switching tubes, compensation capacitors, wires used for     coils, etc. -   6) The primary input current should lag the input square wave to     achieve zero-voltage turn-on. FIG. 4 shows the relationship between     voltage and current, where v_(in) is the output square wave of the     full-bridge inverter after phase shift control, and v₁ is the     fundamental wave of this square wave. i₁ is the input current. To     achieve zero-voltage turn-on, the zero-crossing point of i₁ should     lag behind the zero-crossing point of v_(in), i.e., ϕ<0, that is:

$\begin{matrix} {{{\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1}}{2{\sqrt{2} \cdot V_{DC}}} \right)}} < 0};} & (8) \end{matrix}$ where θ is the phase angle of the input current relative to v₁; V₁ is the effective value of the fundamental wave of the inverter output square wave, which can be calculated by the following equation:

$\begin{matrix} {{V_{1} = {\frac{2\sqrt{2}}{\pi}V_{DC}{\sin\left( \frac{\alpha}{2} \right)}}};} & (9) \end{matrix}$ α is the angle occupied by half wave of the square wave, and the angle is 180° when the duty ratio is 1.

By solving the above equations, the range of C₁ and C₂ that satisfy all the above conditions can be obtained.

The application also provides several specific methods for obtaining the combination of C₁ and C₂ that meet the requirements more quickly. The method of solving a group of inequalities and equations can be used to solve the combination of C₁ and C₂ for the two extreme positions of the strongest coupling and the weakest coupling; traversal calculation can also be used to quickly find all capacitor combinations that meet the conditions.

In the present application, by finding a suitable combination of the primary side capacitance C₁ and the secondary side capacitance C₂, the system can output the required rated power in a larger coupling range under the condition of incomplete compensation.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to explain the technical solutions in the embodiments of the present application or the prior art more clearly, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings in the following description are only examples of the application. For those skilled in the art, other drawings can be obtained based on the provided drawings without creative work.

FIG. 1 is a simplified circuit model of an SS-compensated wireless power transfer system;

FIG. 2 is a finite element simulation model of the coil;

FIG. 3 is the curve of the output power of the SS compensation system changed with the air gap distance;

FIG. 4 is a schematic diagram of the input current lagging behind the input square wave;

FIG. 5 is the variation of the input voltage, input current, the phase difference between the zero-crossing point of the input current and the zero-crossing point of the input square wave, the output power, and the coupling coil transmission efficiency (corresponding to the vertical axis of each column from left to right) with the operating frequency in the range of 10 cm to 20 cm air gap (from top to bottom each line corresponds to an air gap) of the system.

DETAILED DESCRIPTION

In the following, the technical solutions in the embodiments of the present application will be clearly and completely described with reference to the drawings in the embodiments of the present application. Obviously, the described embodiments are only a part of the embodiments of the present application, but not all of the embodiments. Based on the embodiments in the present application, all other embodiments obtained by those skilled in the art without creative efforts shall fall within the protection scope of the present application.

The incompletely compensated wireless transmission and charging system of the present application is based on the SS compensated wireless power transfer system topology, taking a primary side capacitance C₁ and a secondary side capacitance C₂ to specific values, and combining the phase-shift and frequency-modulation control method to meet the given coupling range and given restriction conditions, thereby increasing the coupling range of the system's output rated power;

For any given f, M₁₂, C₁, C₂, assuming that the duty cycle of the inverter output square wave is 1, and the peak value is V_(DC), the effective value of the fundamental wave of the square wave is:

$\begin{matrix} {V_{1} = {\frac{2\sqrt{2}}{\pi}V_{DC}}} & (10) \end{matrix}$

For a given rated output voltage V_(out) and power P_(out), the equivalent alternating current load impedance R_(L) can be calculated from:

$\begin{matrix} {R_{L} = {\frac{8}{\pi^{2}}\frac{V_{out}^{2}}{P_{out}}}} & (11) \end{matrix}$

Assuming that the phase of v₁ is zero, that is, V1=V₁. Substituting V1 and R_(L) into equations (1) and (2), for values of any set of C₁, C₂, M₁₂, f, the currents of the primary and secondary side can be solved by equations (1) and (2), assuming that their effective values are respectively I₁ and I2, and the phase angle of the primary current I₁ is θ. The output power when the input voltage is V₁ can be calculated by the following equation: P_(out_V1)=I₂ ²R_(L)  (12) If P_(out_V1)<P_(out)  (13) it means that the system cannot output the required power under this set of parameters; if P_(out_V1)≥P_(out)  (14) it means that the system can output the required power through phase shift control. At this time, the effective value of the fundamental wave of the square wave required by the system is:

$\begin{matrix} {V_{1{\_{Pout}}} = {V_{1}\sqrt{\frac{P_{out}}{P_{{out\_ V}1}}}}} & (15) \end{matrix}$

The effective value of the input current required by the system can also be obtained:

$\begin{matrix} {I_{1{\_{Pout}}} = {I_{1}\sqrt{\frac{P_{out}}{P_{{out\_ V}1}}}}} & (16) \end{matrix}$

From (9), the phase difference between the current zero-crossing point and the zero-crossing point of the input square wave can be calculated as:

$\begin{matrix} {\phi = {\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1{\_{Pout}}}}{2{\sqrt{2} \cdot V_{DC}}} \right)}}} & (17) \end{matrix}$

Theoretically, it is possible to solve the equations by mathematical methods to obtain the range of the primary and secondary side capacitances C₁ and C₂ that meet all the restrictions, that is, the combination of C₁ and C₂.

In actual operation, based on the above ideas, it is also possible to find all the combinations of C₁ and C₂ that meet the conditions faster by directly using traversal calculations, as follows:

-   a) for a given relative position range of the coil, first use the     finite element simulation method to obtain corresponding coil     self-inductance and mutual-inductance ranges [L_(imin), L_(imax)],     [M_(12min), M_(12max)]; -   b) assuming that the operating frequency of the system is f_(min) at     M_(12min) and that the operating frequency of the system is f_(max)     at M_(12max). To improve the system transfer efficiency at the     weakest coupling position, C′₁, C′₂ under full compensation at     M_(12min) may be calculated; taking C′₁, C′₂ as the center,     determining the range of C₁ and C₂ to undergo traversal: [C_(1min),     C_(1max)], [C_(2min), C_(2max)], the initial range can be set by     oneself, usually 0.5 C′˜1.5 C′. Then, according to a certain step,     taking the capacitor combination value in an exhaustive manner, and     when the calculated operating distances are M_(12min) and M_(12max)     respectively, verifying one by one whether the selected capacitance     combination can meet all the restriction conditions, if it meets,     then keeping the capacitance combination, if it does not meet, then     removing it; -   c) assuming that the operating frequency of the system is f_(max) at     M_(12min) and that the operating frequency of the system is f_(min)     at M_(12min). To improve the system transfer efficiency at the     weakest coupling position, C″₁, C″₂ under full compensation at     M_(12min) may be calculated; taking C″₁, C″₂ as the centers,     determining the range of C₁ and C₂ to undergo traversal: [C_(1min),     C_(1max)], [C_(2min), C_(2max)]. Similarly, the initial range can be     set by oneself, usually 0.5 C″˜1.5 C″. Taking the capacitor     combination value in an exhaustive manner, and when the calculated     operating distances are M_(12min) and M_(12max) respectively,     verifying one by one whether the selected capacitor combination can     meet all the restriction conditions, if it meets, then keeping the     capacitor combination, if it does not meet, then removing it; -   d) the solution set of the capacitor combination that meets the     conditions can be obtained through the above two steps; if the     solution is an empty set, it means that the set restriction     conditions are not reasonable, and the restriction conditions should     be appropriately relaxed, such as increasing the limit of the     primary current, or reducing the air gap range. The capacitor     combination solution that meets the conditions are found through     traversal again.

The above steps find a capacitor combination that can meet the restrictions at the two extreme positions corresponding to M_(12min) and M_(12max). Then taking the air gap distance within [M_(12min), M_(12max)] in a certain step in an exhaustive manner, and obtaining the self-inductance and mutual-inductance data by simulation, and calculating the input voltage, input current, the phase difference between the zero-crossing point of input current and the zero-crossing point of input square wave (that is, ϕ), and the change characteristics of the output power with the operating frequency. Therefore, a system control strategy is formulated by combining phase shift frequency modulation.

Take the wireless power transfer system defined in Table 1 and FIG. 2 as an example. V_(DC) is 400 V; V_(out) is 400 V; P_(out) is 3.3 kW; I_(1max) is 12.6 A; air gap range is 11 cm to 20 cm (coils directly opposite).

First, the step size for traversing the capacitance combination is set to 0.01 pF. Through the above methods, the combinations of C₁ and C₂ that meet the restrictions are found: (6.03 pF, 5.65 pF), (6.04 pF, 5.65 pF), (6.05 pF, 5.65 pF). Because the three combined values are close, the impact on system performance is small. Take (6.03 pF, 5.65 pF) here for verification. In some cases, a relatively large range of capacitance combinations may be obtained. In practice, the error range of the capacitance needs to be considered, and then a combination with high system efficiency and small input current (which affects the efficiency of the inverter and the selection of the switch) is selected. This case study does not consider the error range of the capacitance.

FIG. 5 lists in the range of 10 cm to 20 cm air gap, the change of the input voltage, input current, phase difference (that is ϕ) between the zero-crossing point of the input current and the zero-crossing point of the input square wave, the output power, and the coupling coil transmission efficiency with the operating frequency of the system. Seen from left to right, the vertical axis of the first column is the effective value V1 of the fundamental wave of the input square wave. When the duty cycle is 1, V1 is about 360 V. If the output power is greater than the rated output power when the duty ratio is 1, then V1 is reduced through the phase shift control to make the output power up to 3.3 kW. The vertical axis of the second column is the effective value of the input current. The vertical axis of the third column is the phase difference (i.e., ϕ) between the zero crossing point of the input current and the zero crossing point of the input square wave. The vertical axis of the fourth column is the output power. The fifth column of the vertical axis is the transmission efficiency of the coupling coil. From top to bottom, the corresponding air gap ranges for each row of the curve are: 10 cm, 11 cm, 12 cm, 13 cm, 14 cm, 15 cm, 15.5 cm, 16 cm, 16.5 cm, 17 cm, 17.5 cm, 18 cm, 19 cm, 20 cm.

It can be seen from the results in FIG. 5 that the air gap is from 11 cm to 20 cm, that is, within the set range, the system can output the required rated power. The data points marked in the figure are the operating points under the corresponding air gap.

The specific control strategies adopted are:

Set the starting operating frequency (for example, set at 90 kHz), and set the starting square wave duty cycle (for example, set at 0). With the same frequency, gradually increase the duty cycle until the rated power is output. If the rated power is still not output when the duty cycle is 1, the operating frequency is gradually reduced in a certain step until the system can output the rated power.

As can be seen from FIG. 5 , under this control strategy, the system can achieve zero voltage turn-on in all operating areas.

The maximum primary current of the system is 12.55A (corresponding air gap distance is 20 cm). Compared with the fully compensated wireless energy transmission system, which normally operates with an air gap range of 18 cm to 20 cm when the current is increased by 35%, the method of the present application can greatly increase the coupling range of the system output rated power.

The technical solution provided by the present application has been described in detail above. Specific examples are used herein to explain the principle and implementation of the present application. The description of the above embodiments is only used to help understand the method of the present application and its core ideas. It should be noted that, for those skilled in the art, without departing from the principle of the present application, several improvements and modifications can be made to the present invention, and these improvements and modifications also fall within the protection scope of the claims of the present application. 

What is claimed is:
 1. An incompletely compensated wireless power transfer system, comprising: two coils and two compensation capacitors, wherein on basis of an Series-Series (SS) compensated wireless power transfer system topology, a primary side compensation capacitance C₁ and a secondary side compensation capacitance C₂ take specific values, and a phase shift frequency modulation control method is combined, to increase a coupling range of rated output power of the system; values of the primary side compensation capacitance C₁ and the secondary side compensation capacitance C₂ are obtained by a first method described below: 1) circuit equations of a wireless charging transfer system: (R ₁ +jX ₁)I ₁ +jωM ₁₂ I ₂ =V ₁   (1) jωM ₁₂ I ₁+(R ₂ +R _(L) +jX ₂)I ₂=0   (2) where R_(i), i=1 or 2, is parasitic resistance, generally including coil resistance and equivalent series capacitor resistance; X_(i)=ωL_(i)−1/(ωC_(i)), i=1 or 2, is reactance of a harmonic oscillator i; L_(i), C_(i), i=1 or 2, are coil inductance and compensation capacitance, I₁ and I₂ are current phasors of a primary loop and a secondary loop respectively; V₁ is input voltage phasor; M₁₂ is mutual inductance of the two coils; R_(L) is equivalent load resistance; ω=2πf is angular frequency, and f is frequency of a power supply; 2) with rated output direct current (DC) voltage V_(out) and power P_(out), the equivalent load resistance is obtained by: $\begin{matrix} {R_{L} = {\frac{8}{\pi^{2}}\frac{V_{out}}{P_{out}}}} & (3) \end{matrix}$ 3) range of operating frequency is given by: f_(min)≤f≤f_(max)  (4) wherein f_(min) is a minimum value of f and f_(max) is maximum value of f; 4) coupling range of the two coils is obtained by: M_(12min)≤M₁₂≤M_(12max)  (5) wherein M_(12min) is a minimum value of M₁₂ and M_(12max) is a maximum value of M_(12;) 5) based on DC input voltage V_(DC), through a phase shift control, V_(1_r) is an effective value of a fundamental wave of a square wave applied to a primary LC harmonic oscillator is limited by: $\begin{matrix} {V_{1{\_ r}} \leq {\frac{2\sqrt{2}}{\pi}V_{DC}}} & (6) \end{matrix}$ 6) maximum primary current I_(1max), is obtained by: I₁≤I_(1max)  (7) where, I₁ is an effective value of the current phasor I₁; 7) a primary input current configured to lag behind an input square wave to achieve zero voltage turn-on, is obtained by: $\begin{matrix} {{\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1}}{2{\sqrt{2} \cdot V_{DC}}} \right)}} < 0} & (8) \end{matrix}$ where θ is a phase angle of the primary input current phasor I₁ with respect to an input voltage phasor V₁, and V₁ is an effective value of a fundamental wave of an inverter output square wave: $\begin{matrix} {{V_{1} = {\frac{2\sqrt{2}}{\pi}V_{DC}{\sin\left( \frac{\alpha}{2} \right)}}};} & (9) \end{matrix}$ α is an angle occupied by a half wave of the inverter output square wave, and is 180° when a duty cycle of the inverter output square wave is 1; 8) calculating the coil resistance via a finite element simulation; the equivalent series resistance of a compensation capacitor configured to be estimated by the following method: the coil inductance is L_(i), then the compensation capacitance C_(i) is approximately resonance capacitance value C_(i0), and a calculation method of the resonance capacitance value and corresponding equivalent series resistance is as follows: $\begin{matrix} {{C_{i0} = \frac{1}{\omega^{2}L_{1}}};} & (10) \end{matrix}$ $\begin{matrix} {{R_{Ci} \approx \frac{\tan\delta}{\omega C_{i0}}};} & (11) \end{matrix}$ where tan δ is a tangent value of a loss angle of the compensation capacitor; when in simplified calculation, the parasitic resistance is neglected; for given coils L₁ and L₂, in combination with above operating conditions, ranges of C₁ and C₂ that simultaneously meet conditions is obtained by a numerical solution method.
 2. The incompletely compensated wireless power transfer system according to claim 1, wherein all combinations of C₁ and C₂ that meet the conditions are obtained faster than the first method by solving a group of equations and inequalities, as follows: 1) giving the DC input voltage V_(DC), the rated output DC voltage V_(out) and the power P_(out), the equivalent load resistance is obtained by: $\begin{matrix} {R_{L} = {\frac{8}{\pi^{2}}\frac{V_{out}}{P_{out}}}} & (12) \end{matrix}$ 2) for a given relative position range of a coil of the two coils, obtaining corresponding ranges of the coil inductance and mutual inductance [L_(imin), L_(imax)] (i=1 or 2), [M_(12min), M_(12max)] by the finite element simulation, and calculating the coil resistance ; according to equations (10) and (11), estimating the capacitor resistance at a strongest coupling position or a weakest coupling position, when in the simplified calculation, the parasitic resistance is neglected; 3) at the weakest coupling position, wherein M₁₂=M_(12min), the operating frequency is the highest, so that f=f_(max); when a duty cycle of the input square wave is 1, so that $\begin{matrix} {V_{1} = {\frac{2\sqrt{2}}{\pi}V_{DC}}} & (13) \end{matrix}$ using the input voltage as a reference, so that V₁=V₁; substituting R_(L), M_(12min), L_(1min), L_(2min), f_(max), R₁, R₂ and V₁ into equations (1) and (2), and obtained that: $\begin{matrix} {I_{1} = \frac{V_{1}}{R_{1} + {j\left( {{\omega_{\max}L_{1\min}} - \frac{1}{\omega_{\max}C_{1}}} \right)} + \frac{\omega_{\max}^{2}M_{12\min}^{2}}{\begin{matrix} {R_{2} + R_{L} +} \\ {j\left( {{\omega_{\max}L_{2\min}} - \frac{1}{\omega_{\max}C_{2}}} \right)} \end{matrix}}}} & (14) \end{matrix}$ $\begin{matrix} {I_{2} = \frac{{- j}\omega_{\max}M_{12\min}V_{1}}{\begin{matrix} {{\omega_{\max}^{2}M_{12\min}^{2}} + \left\lbrack {R_{1} + {j\left( {{\omega_{\max}L_{1\min}} - \frac{1}{\omega_{\max}C_{1}}} \right)}} \right\rbrack} \\ \left\lbrack {R_{2} + R_{L} + {j\left( {{\omega_{\max}L_{2\min}} - \frac{1}{\omega_{\max}C_{2}}} \right)}} \right\rbrack \end{matrix}}} & (15) \end{matrix}$ where ω_(max)=2πf_(max;) the output power corresponding to the input voltage V₁ is calculated by the following equation: P_(out_V) ₁ =|I₂|²R_(L)   (16) with the rated output power being P_(out), obtaining the following inequality: P_(out_V) ₁ ≥P_(out)   (17) under equation (17), the system is configured to output a required rated power by the phase shift control; and an effective value of a fundamental wave of a square wave required by the system is: $\begin{matrix} {V_{1{\_ P}_{out}} = {V_{1}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (18) \end{matrix}$ an effective value of an input current required by the system: $\begin{matrix} {I_{1{\_ P}_{out}} = {{❘I_{1}❘}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (19) \end{matrix}$ with a maximum value of a given input current being I_(1max), obtaining the following inequality: I_(1_P) _(out) ≤I_(1max)   (20) a phase difference between a zero crossing point of the input current and a zero crossing point of the input square wave is: $\begin{matrix} {\phi = {\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1{\_{Pout}}}}{2\sqrt{2 \cdot}V_{DC}} \right)}}} & (21) \end{matrix}$ where θ is the phase angle of the primary input current phasor I₁, in order to achieve zero-voltage turn-on of an input inverter, the phase difference is configured to satisfy the following inequality: ϕ<0  (22) 4) at the strongest coupling position, wherein M₁₂=M_(12max), the operating frequency is the lowest, so that f=f_(min); assuming the duty cycle of the input square wave is 1, using the input voltage as a reference, so that V₁=V₁; substituting R_(L), M_(12max), L_(1max), L_(2max), R₁, R₂, f_(min), and V₁ into equation (1) (2), and obtained that: $\begin{matrix} {I_{1} = \frac{V_{1}}{\begin{matrix} {R_{1} + {j\left( {{\omega_{\min}L_{1\max}} - \frac{1}{\omega_{\min}C_{1}}} \right)} +} \\ \frac{\omega_{\min}^{2}M_{12\max}^{2}}{R_{2} + R_{L} + {j\left( {{\omega_{\min}L_{2\max}} - \frac{1}{\omega_{\min}C_{2}}} \right)}} \end{matrix}}} & (23) \end{matrix}$ $\begin{matrix} {I_{2} = \frac{{- j}\omega_{\min}M_{12\max}V_{1}}{\begin{matrix} {{\omega_{\min}^{2}M_{12\max}^{2}} + \left\lbrack {R_{1} + {j\left( {{\omega_{\min}L_{1\max}} - \frac{1}{\omega_{\min}C_{1}}} \right)}} \right\rbrack} \\ \left\lbrack {R_{2} + R_{L} + {j\left( {{\omega_{\min}L_{2\max}} - \frac{1}{\omega_{\min}C_{2}}} \right)}} \right\rbrack \end{matrix}}} & (24) \end{matrix}$ where ω_(min)=2πf_(min); calculating the output power corresponding to the input voltage V₁ by the following equation: P_(out_V) ₁ =|I₂|²R_(L)   (25) with the rated output power being P_(out), obtaining the following inequality: P_(out_V) ₁ ≥P_(out)   (26) the system is configured to output required power through the phase shift control; and the effective value of the fundamental wave of the square wave required by the system is: $\begin{matrix} {V_{1{\_ P}_{out}} = {V_{1}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (27) \end{matrix}$ obtaining the effective value of the input current required by the system as: $\begin{matrix} {I_{1{\_ P}_{out}} = {{❘I_{1}❘}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (28) \end{matrix}$ with the maximum value of the given input current being I_(1max), obtaining the following inequality: I_(1_P) _(out) ≤I_(1max)   (29) the phase difference between the zero crossing point of the input current and the zero crossing point of the input square wave is: $\begin{matrix} {\phi = {\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1{\_{Pout}}}}{2\sqrt{2 \cdot}V_{DC}} \right)}}} & (30) \end{matrix}$ where θ is the phase angle of the primary input current phasor I_(1;) in order to achieve the zero-voltage turn-on of the input inverter, the phase difference is configured to satisfy the following inequality: ϕ<0   (31) 5) solving required ranges of C₁ and C₂ from simultaneous inequalities (17), (20), (22), (26), (29) and (31).
 3. The incompletely compensated wireless power transfer system according to claim 1, wherein all combinations of C₁ and C₂ that meet the conditions are obtained faster than the first method by solving a second group of equations and inequalities, as follows: 1) giving the DC input voltage V_(DC), the rated output direct current voltage V_(out) and the power P_(out), then $\begin{matrix} {R_{L} = {\frac{8}{\pi^{2}}\frac{V_{out}}{P_{out}}}} & (32) \end{matrix}$ 2) for a given relative position range of the coil of the two coils, obtaining corresponding ranges of the coil inductance and mutual inductance [L_(imin), L_(imax)] (i=1 or 2), [M_(12min) , M_(12max)] via the finite element simulation, and calculating the coil resistance; according to equations (10) and (11), estimate the capacitor resistance at a strongest coupling position or a weakest coupling position, when in the simplified calculation, the parasitic resistance is neglected; 3) at the weakest coupling position, wherein M₁₂=M_(12min), the operating frequency is the lowest, so that f=f_(min); when a duty cycle of the primary input square wave is 1, so that $\begin{matrix} {V_{1} = {\frac{2\sqrt{2}}{\pi}V_{DC}}} & (33) \end{matrix}$ using the input voltage as a reference, so that V₁=V₁; substituting R_(L), M_(12min), L_(1min), L_(2min), f_(max), R₁, R₂ and V₁ into equations (1) and (2), and obtained that: $\begin{matrix} {I_{1} = \frac{V_{1}}{\begin{matrix} {R_{1} + {j\left( {{\omega_{\min}L_{1\min}} - \frac{1}{\omega_{\min}C_{1}}} \right)} +} \\ \frac{\omega_{\min}^{2}M_{12\min}^{2}}{R_{2} + R_{L} + {j\left( {{\omega_{\min}L_{2\min}} - \frac{1}{\omega_{\min}C_{2}}} \right)}} \end{matrix}}} & (34) \end{matrix}$ $\begin{matrix} {I_{2} = \frac{{- j}\omega_{\min}M_{12\min}V_{1}}{\begin{matrix} {{\omega_{\min}^{2}M_{12\min}^{2}} + \left\lbrack {R_{1} + {j\left( {{\omega_{\min}L_{1\min}} - \frac{1}{\omega_{\min}C_{1}}} \right)}} \right\rbrack} \\ \left\lbrack {R_{2} + R_{L} + {j\left( {{\omega_{\min}L_{2\min}} - \frac{1}{\omega_{\min}C_{2}}} \right)}} \right\rbrack \end{matrix}}} & (35) \end{matrix}$ where ω_(min)=2πf_(min;) calculating the output power corresponding to the input voltage V₁ by the following equation: P_(out_V) ₁ =|I₂|²R_(L)   (36) with the rated output power being P_(out), obtaining the following inequality: P_(out_V) ₁ ≥P_(out)   (37) under the inequality (37), the system is caused to output a required rated power by the phase shift control; an effective value of a fundamental wave of a square wave required by the system is: $\begin{matrix} {V_{1{\_{Pout}}} = {V_{1}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (38) \end{matrix}$ the effective value of the input current reauired by the system is: $\begin{matrix} {I_{1{\_ P}_{out}} = {{❘I_{1}❘}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (39) \end{matrix}$ with a maximum value of the given input current being I_(1max), obtaining the following inequality: I_(1_P) _(out) ≤I_(1max)   (40) a phase difference between a zero crossing point of the input current and a zero crossing point of the input square wave is: $\begin{matrix} {\phi = {\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1{\_{Pout}}}}{2\sqrt{2 \cdot}V_{DC}} \right)}}} & (41) \end{matrix}$ where θ is a phase angle of the primary input current phasor I₁, in order to achieve zero-voltage turn-on of an input inverter, the phase difference is configured to satisfy the following inequality: ϕ<0   (42) 4) at the strongest coupling position, wherein M₁₂=M_(12max), the operating frequency is the highest, so that f=F_(max); assuming that when the duty cycle of the input square wave is 1, using the input voltage as a reference, that is, V₁=V₁; substituting R_(L), M_(12max), L_(1max), L_(2max), R₁, R₂, f_(min) and V₁ into equations (1) (2), and obtaining the followings: $\begin{matrix} {I_{1} = \frac{V_{1}}{\begin{matrix} {R_{1} + {j\left( {{\omega_{\max}L_{1\max}} - \frac{1}{\omega_{\max}C_{1}}} \right)} +} \\ \frac{\omega_{\max}^{2}M_{12\max}^{2}}{R_{2} + R_{L} + {j\left( {{\omega_{\max}L_{2\max}} - \frac{1}{\omega_{\max}C_{2}}} \right)}} \end{matrix}}} & (43) \end{matrix}$ $\begin{matrix} {I_{2} = \frac{{- j}\omega_{\max}M_{12\max}V_{1}}{\begin{matrix} {{\omega_{\max}^{2}M_{12\max}^{2}} + \left\lbrack {R_{1} + {j\left( {{\omega_{\max}L_{1\max}} - \frac{1}{\omega_{\max}C_{1}}} \right)}} \right\rbrack} \\ \left\lbrack {R_{2} + R_{L} + {j\left( {{\omega_{\max}L_{2\max}} - \frac{1}{\omega_{\max}C_{2}}} \right)}} \right\rbrack \end{matrix}}} & (44) \end{matrix}$ where ω_(max)=2πf_(max;) calculating the output power corresponding to the input voltage V₁, by the following equation: P_(out_V) ₁ =|I₂|²R_(L)  (45) with the rated output power being P_(out), obtaining the following inequality: P_(out_V) ₁ ≥P_(out)   (46) under equation (46), the system is configured to output the required power by the phase shift control; at this time, the effective value of the fundamental wave of the square wave required by the system is: $\begin{matrix} {V_{1{\_ P}_{out}} = {V_{1}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (47) \end{matrix}$ obtaining the effective value of the input current required by the system: $\begin{matrix} {I_{1{\_ P}_{out}} = {{❘I_{1}❘}\sqrt{\frac{P_{out}}{P_{{out\_ V}_{1}}}}}} & (48) \end{matrix}$ with the maximum value of the given input current being I_(lmax), obtaining the following inequality: I_(1_P) _(out) ≤I_(1max)  (49) the phase difference between the zero crossing point of the input current and the zero crossing point of the input square wave is: $\begin{matrix} {\phi = {\theta + {90{^\circ}} - {\arcsin\left( \frac{\pi V_{1{\_{Pout}}}}{2\sqrt{2 \cdot}V_{DC}} \right)}}} & (50) \end{matrix}$ where θ is the phase angle of the primary input current phasor I₁, in order to achieve the zero-voltage turn-on of the input inverter, the phase difference is configured to satisfy the following inequality: ϕ<0   (51) 5) solving required ranges of C₁ and C₂ from simultaneous inequalities (37), (40), (42), (46), (49), (51).
 4. The incompletely compensated wireless power transfer system according to claim 1, wherein all combinations of C₁ and C₂ that meet the conditions are configured to be obtained faster than the first method by using a traversal calculation method, specifically as follows: a) for a given relative position range of a coil of the two coils, first obtaining corresponding ranges of coil inductance and mutual-inductance [L_(imin), L_(imax)], [M_(12min),M_(12max)] via the finite element simulation; b) when the operating frequency of the system is f_(min) at M_(12min) and that the operating frequency of the system is f_(max) at M_(12max); calculating a center primary side compensation capacitance C′₁ and a center secondary side compensation capacitance C′₂ under full compensation at M_(12min); setting C′₁, C′₂ as centers, determining a range of C₁ and C₂ to be undergo traversal of a range of: [C_(1min), C_(1max)], [C_(2min), C_(2max)], selecting a value of capacitance combination in the range of [C_(1min),C_(1max)], [C_(2min), C_(2max)] in an exhaustive manner, and when the calculated coupling is M_(12min) and M_(12max), respectively, verifying one by one to determine whether the selected value of the capacitance combination meets equations (1)-(9), if the selected value of the capacitance combination meets the equations (1)-(9), then keeping the capacitance combination, if the selected value of the capacitance combination does not meet the equations (1)-(9), then removing the capacitance combination; c) when the operating frequency of the system is f_(max) at M_(12min) and that the operating frequency of the system is f_(min) at M_(12max), calculating the center primary side compensation capacitance C″₁ and the center secondary side compensation capacitance C″₂ under full compensation at M_(12min); setting C″₁, C″₂ as the centers, determining the range of C₁ and C₂ to undergo traversal of the range of: [C_(1min), C_(1max)], [C_(2min), C_(2max)], similarly, selecting the value of the capacitance combination in the range of [C_(1min),C_(1max)], [C_(2min), C_(2max)] in the exhaustive manner, and when the calculated coupling is M_(12min) and M_(12max), respectively, verifying one by one to determine whether the selected value of the capacitance combination meets the equations (1)-(9), if the selected value of the capacitance combination meets the equation (1)-(9), then keeping the capacitance combination, if the selected value of the capacitance combination does not meet the equations (1)-(9), then removing the capacitance combination; and d) obtaining a solution set of the capacitance combination that meets the conditions through the above two steps; if the solution set is an empty set, appropriately relaxing restriction in the equations (1)-(9), and conducting traversal again to obtain the solution set of the capacitance combination that meets the conditions. 